Time delay device

ABSTRACT

A time delay circuit for accurately generating an output pulse a preselected time after the occurrance of a trigger pulse. The preselected time may be continuously varied from a predetermined minimum value to a predetermined maximum value. This time delay circuit is particularly suitable for use in the area navigation system disclosed herein.

ilnited States atent Hirsch 1 Aug. 29, 1972 [54] TIME DELAY DEVICE 3,105,939 10/1963 Onno et al. ..328/34 2 h Hirsc P t 3,364,441 1/ 1968 Rogers ..307/293 X [72] Invent C mes J J 3,365,586 1/1968 Billings ..307/293 x [73] Assignee: RCA Corporation 3,469,112 9/1969 Hands et a1 ..330/30 X 3,484,624 12/ 1969 Rasiel et a1. ..328/ 146 X [22] March 1970 3,073,972 l/1963 Jenkins ..307/266 x [21] A L N 24,464 3,277,311 10/1966 Merlen et al. ..307/234 Related US. Application Data Primary Examiner-Donald D. Forrer Assistant ExaminerR. C. Woodbridge [62] I Ell/18;?" :fgifgggo. 746,600, July 22, 1968, Att0mey E-dward J Norton [57] ABSTRACT [52] U.S. Cl. ..307/293, 307/235, 328/146,

328/150 330/30 D A time delay circuit for accurately generating an out- [51] Int Cl 03k 17/28 put pulse a preselected time after the occurrance of a gg pulse. The preselected time y be comimb [58] held of Search ously varied from a predetermined minimum value to 307/267 297; 328/146 a predetermined maximum value. This time delay cir- 330/30 cuit is particularly suitable for use in the area navigaf ed tion system disclosed herein. 56 Re erences Cit 1 3 Claims, 6 Drawing Figures UNITED STATES PATENTS 3,358,218 12/1967 Halpin ..307/228 X f/Z fi F901 a /5 4545 .sW/rcv/ MUL 7710519470,? Mil/VJ 06 1 510 zly i 6 76 V cam/wispy 4:. M54 5 5-: [/8 4 7 {623 .67! 55 i066 a 0514) M54445 mat/arms: 7w: i/Z 410 PATENTEDnuszs m2 SHEET 3 OF 4 :ITYOINIY TIME DELAY DEVICE This invention, which is a division of application Ser. No. 746,600, filed July 22, 1968, now US. Pat. No. 3,534,399, relates to a time delay device and, more particularly, to a time delay device particularly suitable for use in an improved area navigation computer for an aircraft equipped with a VOR and a DME.

VHF-Omnirange (VOR) is one of the most useful and widespread navigation devices. It supplies the azimuth in degrees from North of an aircraft with respect to a preselected one of a plurality of transmitting ground stations, utilized for both VOR and DME, which are distributed over the country at known predetermined locations.

For VOR purposes, each ground station is assigned a given 50 KHz channel within the 108.0 to 117.9 MHz frequency band and emits two RF signals within its assigned channel which are separated by 9,960 hertz in frequency. One of these signals, which is called the reference-phase signal, is frequency modulated at 30 hertz and radiated by an omnidirectional antenna, so that it is received with the same phase at all points which are equi-distant from the antenna. The other of these signals which is called the variable-phase" signal, is not modulated at the transmitter, but is radiated from a directional antenna with a cardioid pattern which rotates at the rate of 30 revolutions per second. This directional antenna is concentric with the omnidirectional antenna which radiates the reference-phase signal. The received variablephase signal, at any point in space, will be amplitudemodulated at 30 hertz because of the rotation of the cardioid pattern at 30 revolutions per second. The phase of this 30 hertz amplitudemodulated signal at any azimuth will differ from the phase thereof at any other azimuth by the difference in the azimuth of the two places. The 30 hertz frequency modulation of the reference-phase signal is synchronized with the 30 r.p.s. of the cardioid pattern of the variable signal so that the received 30 hertz amplitude-modulated variable-phase signal will coincide with the received 30 hertz frequency-modulated reference-phase signal solely in the due North direction with respect to the transmitting ground station. In any other direction, these two signals will differ in phase by an amount equal to the azimuth of that direction with respect to the transmitting ground station.

A VOR receiver aboard an aircraft receives the two RF signals transmitted from a particular ground station to which it is tuned and demodulates these two RF signals to provide at its output a first 30 hertz signal having a phase which manifests the reference-phase and a second 30 hertz signal having a phase which manifests the variable-phase.

Another one of the most useful and widespread navigation devices for an aircraft is distance measuring equipment (DME). Briefly, DME aboard an aircraft, an example of which is described in detail in US. Pat. No. 3,320,612, issued to R. P. Crow et al. on May 16, 1967, transmits an interrogation pulse signal at an appropriate carrier frequency to the same preselected one of the ground stations used for transmitting the received VOR information. Transponder equipment located at this ground station transmits answering pulse signals in response to received interrogation pulse signals. The DME aboard an aircraft measures the then-existing distance between that aircraft and the ground station from which it is receiving answering pulses in accordance with the length of the time interval between the transmission of an interrogation pulse signal therefrom and the receipt thereby of an answering pulse signal from the ground station. The measured distance is indicated at the DME by means of a shaft position which controls the setting of a mechanical counter.

It is often desired to navigate an aircraft equipped with a VOR and a DME with respect to a given position, other than the ground station to which the aircrafts VOR and DME is tuned, which has predetermined known fixed polar coordinates with respect to that ground station. More particularly, since the polar coordinates of the given position with respect to the ground station are known a priori and the then-existing polar coordinates of the aircraft with respect to the ground station are measured by means of the VOR and DME aboard the aircraft, it is possible by trigonometry to provide a transformation of coordinates by which the polar coordinates of the given position with respect to the then-existing position of the aircraft are determined. In order to accomplish this, it is necessary to provide aboard the aircraft area navigation computer means for continuously making the required coordinate transformation in response to the application thereto of the respective reference-phase" and variable-phase" output signals from the VOR and the output from the DME, and by the setting therein of the predetermined known fixed polar coordinates of the given position with respect to the ground station. Further, it is desirable that the computer means include adjustable means for taking care of the case where the course line of the desired flight path is offset a specified amount from the given position.

In the past, computer means for solving the trigonometry involved in the above-discussed area navigation have been relatively complex and costly. This relatively high cost made it impractical to utilize area navigation to any great extent, especially with smaller general aircraft. The present invention is directed to relatively simple and inexpensive computer means required for an area navigation system aboard an aircraft that incorporates a VOR receiver and a DME which makes it practical to extend the benefits of area navigation to nearly all aircraft and the method of utilizing such computer means.

Briefly, the present invention makes use of attenuating, phase-shifting and sampling of the 30 hertz output signals from the VOR receiver in accordance with the azimuth and distance of the given position with respect to the ground station and in accordance with the thenexisting distance of the aircraft from the ground station as measured by the DME in a manner such as to eliminate the sinusoidal time-varying properties exhibited by these VOR output signals, while utilizing the relative phase information contained therein to solve certain predetermined trigonometric functions.

Therefore, it is an object of the present invention to provide an improved aircraft area navigation method and apparatus.

This and other objects, features and advantages of the present invention will become more apparent from the following detailed description taken together with the accompanying drawings in which:

FIG. 1 is a coordinate plot showing the trigonometric relationship among the then-existing position of an aircraft, the position of a ground station and a predetermined known fixed given position;

FIG. 2 is a illustrative block diagram of one embodiment of an area navigating computer means employing the present invention;

FIG. 3 shows a plurality of signal function graphs which are helpful in understanding the operation of the embodiment of the present invention shown in FIG. 2;

FIG. 4 is a block diagram of a modification of the embodiment of the present invention shown in FIG. 2;

FIG. 5 is a block and schematic diagram of an inexpensive variable time delay means particularly suitable for providing an adjustable phase shift in the modification of FIG. 4; and

FIG. 6 shows a plurality of signal function graphs which are helpful in understanding the operation of the embodiment of the present invention shown in FIGS. 4 and 5.

Referring now to FIG. 1, GS. represents the location of a ground station, which is located at the origin of the coordinate plot defined by the N or North coordinate and E or East coordinate. Point A represents the actual position at a given time of an aircraft having a VOR receiver and a DME on board with respect to the ground station. As shown, the azimuth relative to North of aircraft A with respect to the ground station is 0. The radial distance of aircraft A with respect to the ground station is p. Point K represents the predetermined known fixed given position with respect to the ground station. As shown, the azimuth relative to North of given position K with respect to the ground station is 4). The radial distance of given position K from the ground station is r. Further, the azimuth relative to North of the given position K with respect to aircraft A is a and the radial distance of the given position K from aircraft A is d.

From the coordinate plot of FIG. 1, it can be seen that the following trigonometric equations hold true:

In Equations (1) and (2), the values r and d) are the known polar coordinates of the given position K, while 0 is manifested by the phase angle between the variable-phase and reference-phase output signals of the VOR and p is manifested by the output of the DME. Therefore, all the independent variables except a in Equations (1) and (2) are immediately available to an area navigation computer means for solving Equations (1) and (2). However, if a is made an adjustable parameter of such computer means, and is then adjusted to that given value thereof which makes the two different expressions for l in Equation 1-) equal to each other, this given value to which a is adjusted will be the proper value of a to provide the correct solution for Equation (2), thereby ascertaining the value of d. Such a computer means is illustrated in each of FIGS. 2 and 4.

Referring now to FIG. 2, a first signal output from the VOR receiver, which is a 30 hertz sine wave having the reference phase sin wt and the amplitude E is applied across potentiometer 204 and is also applied across potentiometer 206. The wiper of potentiometer 204 is set to a value proportional to distance r and the wiper of potentiometer 206 is automatically set by being mechanically linked to the output shaft of the DME to a value proportional to the distance p.

The signal appearing at the wiper of potentiometer 204, described by rE sin (at, which is shown in curve (1) of FIG. 3a, is applied as an input to sampling means 210 and is also applied as an input to quadrature phase shifter 212. The output of phase shifter 212, which is shown in curve (2) of FIG. 3a, is applied as an input to sampling means 214. It is described by rE cos out. The reference phase signal E sin wt is applied as an input to adjustable phase-shifter 208, which provides a phase shift equal to (b, the bearing of point K. This signal is now equal to E, sin (wt and is shown as curve (1) of FIG. 3b. This signal is then applied to adjustable phase shifter 200 which provides a phase shift of (a). The output of this second adjustable phase shifter is shown as curve (2) of FIG. 3b and is described by E, sin (mt d) a). The output of 200 is then applied to the reference-phase signal sampling pulse generator 202.

Reference-phase signal sampling pulse generator 202 includes a limiter and differentiator circuit for producing a first sampling pulse output, shown in FIG. 3c, in response to each positive-going zero crossover of the modified reference phase first signal applied as an input thereto. The first sampling pulse output from generator 202 is applied as a control signal to each of sampling means 210 and 214 to effect the sampling of the instantaneous level of the sine wave signals then being applied thereto from phase shifters 208 and 212, respectively.

Each of sampling means 210 and 214 includes normally open switch means for sampling the instantaneous level of the input applied thereto in response to the closing of the switch means and, preferably, also includes means for storing the same level until the next sampling. For instance, the switching means of each of sampling means 210 and 214 may comprise a conventional diode quad and the storing means may include a conventional integrating operational amplifier. Another example of a switching means and storing means that can be made inexpensively as an integrated circuit is an MOS FET circuit which takes advantage of the extremely high input impedance of MOS field effect transistors in their cutoff condition for storing the sampled signal between samplings.

The output appearing on the wiper of potentiometer 206, which is shown in curve (1) of FIG. 3d, is applied as an input to sampling means 216 and is also applied as an input to quadrature phase shifter 218. The output from phase shifter 218, which is shown in curve (2) of FIG. 3d, is applied as an input to sampling means 220. Sampling means 216 and 220 are identical in structure and function to sampling means 210 and 214 described above.

The variable phase second output signal from the VOR, which is designated by the function H sin (mt 0) shown in curve (1) of FIG. 30, is applied as an input to adjustable phase shifter 200a to shift the variable phase signal by a. Adjustable phase shifter 200a is mechanically coupled to identical adjustable phase shifter 200 to provide both adjustments simultaneously. The output of phase shifter 200a is then E sin (wt 0 a) and is shown in curve (2) of FIG. 30. It is then applied to variable-phase signal sampling pulse generator 222. Generator 222 includes a limiter and differentiator circuit for producing a second sampling pulse output shown in FIG. 3f, in response to each positive going zero crossover of the variable phase signal applied as an input thereto. The second sampling pulse output from generator 222 is applied as a control signal to each of sampling means 216 and 220 to effect the sampling of the instantaneous level of the sine wave signals then being applied thereto from potentiometer 206 and phase shifter 218, respectively.

The output sample voltage of sampling means 210, which is derived across load resistance 224, is a DC signal having a level and polarity indicative of the value rE sin (4, a). The output sample voltage of sampling means 214, which is derived across lead resistance 226 is a DC signal having a level and polarity indicative of the value rE, cos (d) a). The output sample voltage of sampling means 216, which is.derived across load resistance 228, is a DC signal having a level and polarity indicative of the value pE sin (6 a). The output sample voltage of sampling means 220, which is derived across load resistance 230, is a DC signal having a level and polarity indicative of the value pE cos a).

Distance motor means 232 is connected as shown between load resistance 226 of sampling 214 and load resistance 230 of sampling means 220, whereby distance motor means 232 will indicate the difference between the output sample voltage of sampling means 214 and the output sample voltage of samplingmeans 220. With switch 234 in its a switch position, course deviation indicator 236 is connected as shown directly between load resistance 224 of sampling means 210 and load resistance 228 of sampling means 216, whereby course deviation indicator 236 indicates the difference between the output sample voltage of sampling means 210 and the output sample voltage of sampling means 216.

FIG. 2 further includes means for biasing course deviation indicator 236 when switch 234 is in its b switch position. This biasing means consists of potentiometer 238 connected across two serially connected voltage sources each having a value of E As shown, the midpoint of potentiometer 238 and the midpoint of the voltage source are both connected to the output of sampling means 216 appearing across load resistance 228. Switch 234, when in its b position, is effective in connecting course deviation indicator 236 between load resistance 224 of sampling means 210 and the adjustable wiper of potentiometer 238, whereby course deviation indicator 236 will indicate the difference between the output sample voltage appearing across load resistance 224 and the voltage appearing at the adjustable wiper of potentiometer 238.

Referring now to the operation of the apparatus shown in FIG. 2, switch 234 is initially placed in its a switch position, adjustable phase shifter 208 is adjusted to the known azimuth d) of the given position with respect to the ground station. Potentiometer 204 is adjusted to a value proportional to the known distance r between the given position and the ground station, and potentiometer 206 is automatically adjusted by the shaft output of the DME to which it is mechanically coupled to a value proportional to the distance p between the aircraft and the ground station as measured by the DME. In general, these adjustments to potentiometers 204 and 206 and to phase shifter 208 will not in and of themselves cause the output sample voltage from sampling means 210 appearing across load resistance 224 and the output sample voltage from sampling means 216 appearing across load resistance 228 to be equal to each other. Therefore, course deviation indicator 236 will indicate some other value than zero. However, by manually adjusting adjustable phase shifter 200 to change the value of the phase shift angle a to that given value thereof which causes course deviation indicator 236 to indicate zero, the output sample voltage of sampling means 210 appearing across load resistance 224 may be made equal to the output sample voltage of sampling means 216 appearing across load resistance 228. Thus, this given value of the phase shift angle a is the value thereof which solves trigonometric Equation (I) discussed above in connection with FIG. 1. Further, the distance indicated by distance meter means 232 when phase shift angle a is adjusted to the aforesaid given value thereof is the solu tion of trigonometric Equation (2), discussed abovein connection with FIG. 1.

If after adjustable phase shifter 200 is adjusted to provide that given value of phase shift angle a which causes course deviation indicator 236 to provide a zero indication, the pilot flies a flight path which maintains the zero indication of course deviation indicator 236, the course line of this flight path will be the line determined by points A and K of FIG. 1. As the aircraft flies along this course line, distance meter means 232 will continuously indicate the correct then-existing distance between the aircraft and the given position.

Therefore with switch 234 in its a switch position the course line of the flight path of the aircraft must intersect the given position Kin FIG. 1. However, it is often desired to navigate the aircraft with respect to the given position K, whose polar coordinates with respect to the ground station are fixed, known and predetermined, while flying a flight path whose course line is spaced a desired distance from this given position K of FIG. .1, rather than intersecting this given position. In this latter case, after adjustable phase shifter 200 is adjusted to provide that given value of phase shift angle a to cause course deviation indicator 236 to provide a zero indication with switch 234 in its a switch position, switch 234 is switched to its b switch position to introduce the biasing means into the circuit with course deviation indicator 236. With switch 234 in its b switch position, the adjustable wiper of potentiometer 238 is offset from the midpoint thereof in a direction and by an amount determined by the location of the desired offset course line of the flight path with respect to given position K of FIG. 1. This will result in course deviation indicator 236 indicating some certaindeviation, either to the left or right as the case may be, from zero. The pilot can then reach his desired offset course line by flying the aircraft in that direction, to the left or right as the case may be, which brings the indication of course deviation indicator 236 back to zero. After the aircraft has reached its desired oflset course line, its flight path may be kept thereon by piloting the aircraft to maintain course deviation indicator 236 at its zero indication. This will result in the flight path of the aircraft having an offset course line which is spaced from but is parallel to the line determined by point A and point K of FIG. 1.

In this case, after the desired offset course line is reached, so that course deviation indicator 236 indicates zero, distance meter means 232 will continuously indicate the correct distance between the thenexisting position of the aircraft and the projection of the given position point K on the desired oflset course line.

Referring now to FIG. 4, there is shown a modification of the apparatus of FIG. 2. In FIG. 4, elements 204, 206, 210, 212, 214, 216, 218, and 220 are identical in structure and function to their correspondingly numbered elements of FIG. 2. In addition, although not specifically shown in FIG. 4, the outputs of sampling means 210, 214, 216, and 220 are coupled to a course deviation indicator and a distance meter means in a manner identical to that shown in FIG. 2.

However, as opposed to FIG. 2, in FIG. 4 adjustable phase shifters 200 and 208 and reference phase signal sampling pulse generator 202 are replaced by phase shifter 400, monostable multivibrator 402, and first adjustable time delay means 404. Adjustable phase shifter 200a and variable-phase signal sampling pulse generator 222 are replaced by phase shifter 406, monostable multivibrator 408, and second adjustable second time delay means 410.

Phase shifter 400 shifts the phase of the signal output therefrom with respect to the signal input thereto in a negative direction by a fixed certain angle such as 15. The input to phase shifter 400 is shown in FIG. 6a and the output from phase shifter 400 is shown in FIG. 6b. They differ in phase with respect to each other by 15, as shown in FIG. 60.

The output from phase shifter 400 is applied as an input to monostable multivibrator 402, which in response thereto produces a first controlpulse having its leading edge in time coincidence with the zero degree crossover of the output from phase shifter 400. This first control pulse has a fixed certain duration, such as 420, which as shown in FIG. 6d is greater than one and less than two periods of the sinusoidal output signal from the VCR. First control pulses from monostable multivibrator 402 are applied as an input to first adjustable time delay means 404 which, as will be described in more detail below, produces a first sampling pulse as an output therefrom after angular delay (4) a) which can be adjusted from a minimum value equal to the phase shift provided by phase shifter 400 to a maximum value which exceeds this minimum value by at least 360, but which is less than the duration of a first control pulse. First sampling pulses from adjustable time delay means 404 are applied as a control signal to sampling means 210 and 214 in the same manner as previously described in connection with FIG. 2.

The variable phase sinusoidal output signal from the VOR is applied as an input to phase shifter 406, which is identical to phase shifter 400, and produces as an output a sinusoidal wave which is shifted in phase with respect to a signal input thereto in a negative direction by a fixed certain angle, such as 15. The input to phase shifter 400 and the output therefrom will be similar to the waves shown respectively in FIG. 6a and FIG. 6b, except that they will be phase displaced with respect thereto by an amount equal to the azimuth 0 of the aircraft.

The output from phase shifter 406 is applied as an input to monostable multivibrator 408, which in response thereto produces a second control pulse having its leading edge in time coincidence with the zero degree crossover of the output from phase shifter 406.

This second control pulse has, a fixed certain duration, such as 420, which is greater than one and less than two periods of a sinusoidal output from the VCR. Second control pulses from monostable multivibrator 408 are applied as an input to second adjustable time delay means 410, which produces a second sampling pulse as an output therefrom after a time delay a which can be adjusted from a minimum value equal to the phase shift provided by phase shifter 406 to a maximum value which exceeds this minimum value by at least 360, but which is less than the duration of a control pulse from monostable multivibrator 408. Second sampling pulses from adjustable time delay means 410 are applied as a control signal to sampling means 216 and 220 in the same manner as previously described in connection with FIG. 2.

A circuit which is particularly suitable for use as first time delay means 404, and which may utilize integrated circuitry to a large extent is shown in FIG. 5. An electronic switch, which is shown schematically as element 500, of switch means 501 is effective when closed in providing a charging circuit having a short time constant for charging capacitance 502 to the voltage V of DC voltage source 504. Capacitance 502 is provided with a discharge circuit consisting of resistance 506 including a first linearly adjustable resistance portion 506a thereof which may be varied from a minimum value thereof when wiper 508a is adjusted to point 5 l l to a maximum value thereof when wiper 508a is adjusted to point 510. Resistance 506 further includes a second independently linearly adjustable resistance portion 506b serially connected to first linearly adjustable resistance portion 506a which may be varied from a minimum value thereof when wiper 508k is adjusted to point 511 to a maximum value thereof when wiper 508b is adjusted to point 512. However, so long as switch 500 is closed, this discharge circuit is without effect and capacitance 502 remains charged to voltage V. Switch 500 of switch means 501 is normally closed, but is opened during the existence of each first control pulse applied as an input thereto from monostable multivibrator 402. Capacitance 502 discharges through resistance 506 while switch 500 is opened. As will become more apparent later, the leading edge of each first control pulse applied to switch means 501 acts as a first start signal which initiates a first time delay interval.

Connected across voltage source 504 is adjustable voltage divider 516 having a movable tap 518 connected as a first input to comparison means 520. The voltage across capacitance 502 is applied as a second input to comparison means 520. Comparison means 520, which may be a differential amplifier for instance, produces an output pulse in response to the potential at its first and second inputs being equal to each other. The output pulse from comparison means 520 is utilized as the first sampling pulse output of the adjustable first time delay means.

Considering now the operation of the adjustable first time delay means shown in FIG. 5, at the time switch 500 is opened in response to a first control pulse being applied to switch means 501 from monostable multivibrator 402, capacitance 502 has already been charged to its maximum voltage V. Therefore, the potential at the second input of comparison means 520 will then be higher than the potential applied to the first input of comparison means 520 from movable tap 518 of voltage divider 516. However, in response to the opening of switch 500, capacitance 502 begins to discharge through resistance 506 at a rate which is determined by the respective resistance values to which each of serially connected linearly adjustable resistance portions 506a and 506b is then adjusted. When capacitance 502 has discharged to that point where the respective potentials of the first and second inputs to comparison means 520 are equal to each other, an output pulse from comparison means 520 will be applied as a first sampling pulse to sampling means 210 and 214 of FIG. 4. Capacitance 502 will continue to be discharged through resistance 506 until the termination of the first control pulse then being applied to switch means 501. In response to the termination of this first control pulse, however, switch 500 of switch means 501 is closed, causing capacitance 502 to again be charged up to its maximum voltage V before the application of the next occurring first control pulse to switch means 501. I

It will be seen that the time between the occurrence of a first start signal, manifested by the leading edge of a first control pulse, and the next-occurring first sampling pulse obtained from the output of comparison 520 depends both on the setting of movable tap 518 of voltage divider 516 and the respective settings of wipers 508a and 508b of linearly adjustable resistance portion 506a and 506b. It is desirable that the time delay between the first sampling pulse and the first start signal be continuously variable over an interval at least equal to one period of the output signals from the VCR, i.e., one-thirtieth of a second. At the same time it is essential that the minimum resistance to which resistance 506 can be adjusted is sufficiently high as not to short circuit capacitance 502 and thereby prevent capacitance 502 from being charged to voltage V. Therefore, there has to be some minimum time delay between the occurrence of a first start signal and the occurrence of a first sampling pulse in response thereto which is greater than zero. This minimum time delay is made to be just equal and opposite to the phase shift provided by phase shifter 400, i.e., l/720th of a second or of the output sinusoidal signals obtained from the VOR. This is accomplished by adjusting each of wipers 508a and 50812 to provide resistance 506 with its predetermined minimum resistance. Then, movable tap 518 of voltage divider 516 is adjusted to that point at which the time delay between the occurrence of a first start signal manifested by the leading edge of a first control pulse and the occurrence of a first sampling pulse in response thereto is exactly equal to the required minimum time delay.

The maximum value of resistance 506 is chosen to be sufficiently great so that in the case where linearly adjustable resistance portion 506a has its minimum value and linearly adjustable resistance 506b has its maximum value and also in the case where linearly adjustable resistance portion 506a has its maximum value and linearly adjustable resistance portion 506b has its minimum value, the time delay provided will exceed the minimum time delay by at least one period of the output signals obtained from the VCR, one-thirtieth of a second, but is less than the duration of a first control pulse, i.e., 420 in the illustrative example shown in FIG. 6d.

The position of wiper 508a of linearly adjustable resistance portion 506a is manually controlled by a knob which is calibrated in degrees of phase angle over an interval of at least 360 and is utilized for entering the known azimuth of the given position into the computer. The position of wiper 508b of linearly adjustable resistance portion 506b is also manually controlled by a knob which is calibrated in degrees of phase angle over an interval of at least 360 and is utilized for entering an adjustable value of the phase shift angle a into the computer.

FIG. 6e shows the discharging characteristics of capacitance 502 during the presence of a first control pulse and the charging characteristics of capacitance 502 during the absence of a first control pulse or two different settings of the resistance of resistance 506. FIG. 6f shows the relative time of occurrence of a first sampling pulse for each of these two different resistance settings of resistance 506.

With one exception, time delay means 410 is identical in structure to time delay means 404. More particularly, since the time delay provided by time delay means 410 is solely a function of the phase shift angle a, rather than a function of both the given position azimuth iii and the phase shift angle a, as is the case with time delay means 404, resistance portion 5060 of resistance 506 is eliminated in the discharge circuit of time delay means 410. Thus, the resistance 506 of time delay means 410 includes only a single linearly adjustable resistance portion rather than two independent linearly adjustable resistance portions as is the case for time delay means 404. Since the settings of both adjustable resistance portion 506b of time delay means 404 and the adjustable resistance portion of time delay means 410 manifest the value of the phase shift angle a to be entered into the computer, it is preferable that wipers 508b of time delay means 404 and the wiper utilized to control the setting of the adjustable re;- sistance of time delay means 410 be ganged together and manually controlled by a single knob which is calibrated in degrees of phase angle over an interval of at least 360.

While the accuracy of the distance measurement made by distance meter means 232 does depend upon the absolute amplitude E of the reference phase signal, this amplitude will vary by only a few percent above or below a predetermined value from one ground station to another. For most purposes this accuracy of distance measurement is sufficient. However, if even greater accuracy is required, it may be obtained by passing the reference phase output signal from the VCR through a limiter or overdriven amplifier to obtain a 30 hertz square wave having a fixed amplitude independent of any variation in the amplitude of the reference phase output signal from the VOR. This square wave signal can then be passed through a low pass filter to regain a sinusoidal 30 hertz signal having the reference phase and a fixed amplitude which is independent of any 1 l variations in the amplitude of the reference phase output signal from the VCR.

What is claimed is:

1. Time delay means comprising first and second terminals adapted to be connected to a source of fixed voltage, voltage divider resistance means having one end thereof connected directly to said first terminal and the other end thereof connected directly to said second terminal, said voltage divider resistance means including a tap intermediate the ends thereof for deriving therefrom a constant voltage of a given value, a given capacitance, a charging circuit for said capacitance including a trigger-pulse controlled switch means for connecting said entire capacitance across said first and second terminals only when said switch means is in a closed position, a discharging circuit for said capacitance including a linearly adjustable resistance having a predetermined minimum value and a predetermined maximum value connected directly across said capacitance, said switch means switching from a closed position thereof to an open position thereof in response to a start signal being applied thereto, and comparison means having a first input thereof connected directly to said tap (a point) of said voltage divider resistance means (intermediate the ends thereof), said comparison means having a second input thereof connected directly to said charged capacitance for generating an output pulse from said comparison means in response to the potential difference between said first and second inputs thereto (of said comparison means) reaching a predetermined value during the discharge of said capacitance.

2. The time delay means defined in claim 1, wherein said discharging circuit includes first and second serially connected independent linearly adjustable resistances across said capacitance.

3. The time delay means defined in claim 1, wherein said voltage-divider tap is adjustable to set the magnitude of said given value of said constant voltage. 

1. Time delay means comprising first and second terminals adapted to be connected to a source of fixed voltage, voltage divider resistance means having one end thereof connected directly to said first terminal and the other end thereof connected directly to said second terminal, said voltage divider resistance means including a tap intermediate the ends thereof for deriving therefrom a constant voltage of a given value, a given capacitance, a charging circuit for said capacitance including a trigger-pulse controlled switch means for connecting said entire capacitance across said first and second terminals only when said switch means is in a closed position, a discharging circuit for said capacitance including a linearly adjustable resistance having a predetermined minimum value and a predetermined maximum value connected directly across said capacitance, said switch means switching from a closed position thereof to an open position thereof in response to a start signal being applied thereto, and comparison means having a first input thereof connected directly to said tap (a point) of said voltage divider resistance means (intermediate the ends thereof), said comparison means having a second input thereof connected directly to said charged capacitance for generating an output pulse from said comparison means in response to the potential difference between said first and second inputs thereto (of said comparison means) reaching a predetermined value during the discharge of said capacitance.
 2. The time delay means defined in claim 1, wherein said discharging circuit includes first and second serially-connected independent linearly adjustable resistances across said capacitance.
 3. The time delay means defined in claim 1, wherein said voltage-divider tap is adjustable to set the magnitude of said given value of said constant voltage. 